Noise measurement system

ABSTRACT

A method of determining noise in a CATV channel, wherein the CATV channel comprises a predetermined frequency band, that employs digital signal processing to isolate the noise signal from an in-service television signal. The method includes an initial step of obtaining a television signal corresponding to a predetermined CATV channel, the television channel comprising a carrier signal modulated by an information signal. Thereafter, the method includes the step of sampling at least a part of the television signal to produce a digital signal segment, said digital signal segment comprising a carrier component, a noise signals component, and an information signal component, wherein said information signal component has a substantially predetermined signal pattern. The method then employs digital signal processing to separate the carrier component from the digital signal segment to produce a baseband signal substantially comprising the information signal and the noise signal. Finally, an estimate of the information signal is obtained and then subtracted from the baseband signal, thereby producing a noise signal estimate.

This is a continuation of U.S. patent application Ser. No. 09/021,612,filed on Feb. 10, 1998 now U.S. Pat. No. 6,219,095.

FIELD OF THE INVENTION

The present invention relates generally to the field of radio frequencysignal testing, and more particularly, to cable television signaltesting.

BACKGROUND OF THE INVENTION

The radio frequency (“RF”) signals used to transmit information overcable television (“CATV”) distribution systems are subject to severaltypes of undesirable noise. For example, snow noise, composite secondorder (“CSO”) noise, and composite triple beat (“CTB”) noise are threewell-known sources of noise typically found in RF signals transmittedover CATV distribution systems. In efforts to improve service, CATVservice providers perform noise measurements to quantify the performancelevel of the network, or a portion thereof.

Such CATV noise measurements typically include measurements ofparticular types of noise, i.e., CTB or CSO noise, for diagnosis of CATVdistribution system problems. If a particular type of noise isdetermined to be unusually high, while other types of noise aredetermined to be at normal levels, the CATV service provider can morereadily determine the appropriate corrective action to be taken. Forexample, if the CSO noise level is measured to be unusually high, thenthe service provider might try a first set of corrective actions. If,however, the CTB noise is measured to be unusually high, then theservice provider might try a second set of corrective actions. Thus,knowledge of both the presence and type of noise is important in CATVsystem diagnostics.

Moreover, federal regulations require that CATV service providersperform a plurality of noise measurements on a regular basis, including,for example, a CTB noise measurement. CTB noise is caused byconcentrations of triple beat signals that occur near channelfrequencies in cable television systems. As is well known in the art,the triple beat signals are caused by the beating of signals from threeother CATV channels. The resulting CTB causes interference, therebyreducing signal quality. In United States CATV systems, CTB noise isgenerally concentrated within 15 Khz of the visual carrier frequency ofselect channels in a CATV broadband signal. By contrast, CSO noise isconcentrated near 0.75 MHz and 1.25 MHz above the carrier frequency andnear 0.75 MHz and 1.25 MHz below the carrier frequency. Snow noise,moreover, is more or less evenly distributed across the entire CATVbroadband signal spectrum. Because CTB noise is concentrated so close tothe channel frequency, CTB noise is often more difficult to measure thansnow noise or CSO noise.

Because the CTB noise energy is located near the carrier frequency, eachchannel historically had to be taken out of service in order tofacilitate the CTB measurement. Specifically, a technician would firstattach test measurement equipment to a remote site on the cabledistribution network and measure the carrier level of the CATV channelwhile the channel was in-service (or producing a test pattern). Thetechnician would then cause the CATV transmitter to take a channel outof service, or in other words, remove the carrier from that channel.Once the carrier was removed, the technician would perform a signalstrength or power measurement within a band of +/−15 Khz from thecarrier frequency. Because the carrier signal had been removed from thechannel, the measured signal power found within +/−15 Khz of the carrierfrequency constituted the CTB noise level. The ratio of the noise powerlevel to the carrier power level constituted the CTB noise measurement.Once the CTB noise measurement was completed, the CATV channel could beplaced in-service again.

The above described method has at least one severe drawback.Specifically, removing the carrier signal causes interruptions in CATVservice, which is undesirable for several reasons. Interruptions in CATVservice often lead to customer dissatisfaction and customer complaints.

To reduce service interruptions, U.S. Pat. No. 5,617,137 to Whitlowshows an in-service CTB noise measurement system that measures the CTBnoise level of an active television signal. The Whitlow method firstdemodulates the received television signal to produce a basebandtelevision signal including the noise signal. Then, the baseband signalinformation is removed, leaving only the noise signal. An RMS typemeasurement is then performed on the noise signal. To obtain a CTB noisemeasurement, the baseband signal is first filtered such that it containsprimarily only frequencies in the CTB noise energy spectrum.Accordingly, the RMS measurement yields, in theory, on the CTB noise.

One drawback of the Whitlow method described above is the method inwhich the baseband signal information is removed to produce the noisesignal alone. According to the Whitlow method, a portion of one frame ofbaseband signal information is subtracted from the same portion of asubsequent frame of baseband signal information. The portion of thebaseband signal information that is used is a repeating pattern known asthe vertical synchronization interval. The drawback of the Whitlowmethod relates to the subtraction of one frame from a subsequent frameto eliminate the baseband signal information. Specifically, suchsubtraction also subtracts the CTB noise of one frame from the CTB noiseof the subsequent frame. While the CTB noise will often vary greatlyfrom frame to frame, the subtraction of one frame of CTB noise fromanother frame of CTB noise will not produce a reliable CTB noiseestimate.

In addition, the Whitlow method requires analog demodulation devices andfiltering devices which undesirably add to the cost of a device designedto carry out the method. In particular, the Whitlow method requires avideo demodulator, a low pass filter for isolating the CTB noise, and aband pass filter for generating non-CTB noise floor reference values.Such devices not only add to the cost of the device, but they alsooccupy valuable circuit board space, and increase the overall size,weight and energy consumption of the device. The space and energyrequirements are especially important due to the intended portablenature of the device.

Finally, another drawback to the Whitlow method is that CTB noisemeasurements are subject to errors contributed by hum noise. Hum noiseis 60 Hz noise produced by amplifiers in the CATV distribution system.Hum noise results from imperfect isolation of the 60 Hz power linesignal in the power supplies of the amplifiers. It is noted that Whitlowprovides some attenuation of hum noise through its subtraction of onevideo frame from another video frame. Because video frames occur at afrequency of 59.97 times per second, much of the 60 Hz noise is filteredby the subtraction of information in subsequent frames. However, becausehum noise can be substantial compared to CTB noise, even a smalldifference between the frame rate and the hum noise frequency can resultin insufficient filtering to remove hum noise from the CTB noisemeasurement.

As a result, there exists a need for more energy, cost and spaceefficient in-service noise testing method for use in CATV distributionsystems. Moreover, there exists a need for such an in-service noisetesting method that has increased accuracy.

SUMMARY OF THE INVENTION

The present invention fulfills the above-stated needs as well as otherneeds by providing a method and apparatus for performing in-servicenoise measurements in a CATV channel using digital processing methodsfor demodulation and spectral analysis. The use of digital processingmethods for demodulation and spectral analysis replaces analog filterand demodulation circuitry, thereby reducing the size, cost and energyconsumption of the measurement device. In addition, such digitalprocessing methods include generating normalized baseband signalinformation for subtracting from a baseband signal to produce a noisesignal estimate. The subtraction of normalized baseband signalinformation from a measured signal provides a more accurate signal thanprior methods that subtracted one measured noisy signal from anothermeasured noisy signal.

An exemplary method according to the present invention comprises amethod of determining noise in a CATV channel, wherein the CATV channelcomprises a predetermined frequency band, and wherein the CATV channelis considered to be in-service when a transmitted carrier signal havinga frequency in the predetermined frequency band is present. The methodincludes an initial step of obtaining a television signal correspondingto a predetermined CATV channel, the television channel comprising acarrier signal modulated by an information signal. Thereafter, themethod includes the step of sampling at least a part of the televisionsignal to produce a digital signal segment, said digital signal segmentcomprising a carrier component, a noise signal component, and aninformation signal component, wherein said information signal componenthas a substantially predetermined signal pattern. The method thenemploys digital signal processing to separate the carrier component fromthe digital signal segment to produce a baseband signal comprisingsubstantially the information signal component and the noise signalcomponent. Finally, an estimate of the information signal is obtainedand then subtracted from baseband signal, thereby producing a noisesignal estimate.

An exemplary apparatus according to the present invention comprises anapparatus for determining noise in a CATV channel, wherein the CATVchannel comprises a predetermined frequency band and the CATV channel isconsidered to be in-service when a transmitted carrier signal having afrequency in the predetermined frequency band is present. The apparatusincludes an A/D converter and a digital signal processing circuit. TheA/D converter has an input for attachment to a source of televisionsignals corresponding to a select CATV channel, such television signalseach comprising a carrier signal modulated by an information signal. TheA/D converter is operable to sample at least a part of a televisionsignal to produce a digital signal segment, said digital signal segmentcomprising a carrier component, a noise signal component, and aninformation signal component, and wherein the information signal has asubstantially predetermined signal pattern. The digital signalprocessing (“DSP”) circuit is operably connected to receive the digitalsignal segment from the A/D converter. The DSP circuit is operable tofirst separate the carrier component from the digital signal segment toproduce a baseband component, comprising substantially the informationsignal component and the noise signal component. The DSP circuit is thenfurther operable to subtract an estimate of the information signal fromthe baseband signal, thereby producing a noise signal estimate.

The present invention thus provides an in-service noise measurementmethod and apparatus that has sufficient sensitivity to measure CTBnoise, which is located close to each carrier frequency. The presentinvention may also employ digital signal processing techniques to removehum noise from a CTB noise measurement.

The above features and advantages, as well as others, will becomereadily apparent to those of ordinary skill in the art by reference tothe following detailed description and accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a block diagram of a noise measurement circuit according tothe present invention;

FIG. 2 shows a carrier recovery demodulator for use in the noisemeasurement circuit of FIG. 1;

FIG. 3 shows a square law demodulator for use in the noise measurementcircuit of FIG. 1;

FIG. 4A shows a first embodiment of a carrier recovery block for use inthe carrier recovery demodulator of FIG. 2;

FIG. 4B shows a second embodiment of a carrier recovery block for use inthe carrier recovery demodulator of FIG. 2;

FIG. 5A shows an information signal extractor for use in connection withthe noise measurement circuit of FIG. 1 that employs a carrier recoverydemodulator;

FIG. 5B shows an information signal extractor for use in connection withthe noise measurement circuit of FIG. 1 that employs a square lawdemodulator;

FIG. 6 shows an exemplary embodiment of an analog circuit for use in thenoise measurement circuit of FIG. 1; and

FIG. 7 shows a partial frequency response of an exemplary noise signalestimate generated by the noise measurement circuit of FIG. 1.

DETAILED DESCRIPTION

FIG. 1 shows a block diagram of an exemplary noise measurement circuit10 that is operable to receive CATV broadband signals and generate adigital noise signal estimate representative of the noise level of oneof the CATV channels within the CATV broadband signal. The noisemeasurement circuit 10 is further operable to perform digital processingtechniques on the noise signal estimate to obtain a measurement of oneof a plurality of noise types.

The noise measurement circuit 10 includes an analog radio frequency(“RF”) circuit 12, an analog to digital (“A/D”) converter 14, a digitalsignal processing circuit 16, a controller 24, a display 25, and a clock26. The digital signal processing circuit 16 includes a demodulator 18,an information signal extractor 20, a noise processing circuit 22 and anoise calculator 23

The analog circuit 12 comprises an analog receiver and tuner thatreceives a broadband CATV signal, tunes to a select channel and producesa down-converted television signal corresponding to the select channel.FIG. 6 shows an exemplary analog circuit that includes an input stage 52and a tuning/filtering circuit 54. The input stage 52 includes front endcircuitry such as preamplifiers and filters. The tuning/filteringcircuit 54 includes frequency conversion stages and additional filtersand is operable to provide a down-converted television signal centeredat an intermediate frequency (“IF”) and having a predefined bandwidth.To tune to a particular channel, the tuning/filtering circuit 54 of theanalog circuit 12 selects a down conversion frequency such that thecarrier frequency of the channel is converted to the IF.

The IF in the embodiment described herein is 2 MHz. The predefinedbandwidth of the down-converted television signal is preferably ±1.5 MHzfrom the IF in order to include frequencies in which most common typesof identifiable noise are located. If only CTB noise is to be measured,then the predefined bandwidth may suitably be reduced to ±150 kHz. fromthe IF.

Thus, the overall function of the analog circuit 12 is to receive a CATVbroadband signal, typically between 5 and 1000 MHz, and then selectivelydown convert and filter the broadband signal to produce a down-convertedtelevision signal, wherein said down-converted television signalconsists substantially of a down-converted version of a select CATVchannel to be tested or measured. The controller 24 is operablyconnected to the analog circuit 12 to control the channel tuningfunction discussed above.

The resulting down-converted television signal comprises a basebandsignal modulated onto a down-converted carrier frequency. In anexemplary embodiment, the down-converted carrier frequency of IF carrierhas a frequency of 2 MHz.

The analog circuit 12 of FIG. 6 is represented at the functional blocklevel for the purposes of clarity of exposition. Specific analogcircuits capable of performing such tuning and filtering are well knownto those of ordinary skill in the art.

Referring again to FIG. 1, the analog circuit 12 is connected to providethe down-converted television signal to the A/D converter 14. The A/Dconverter 14 may suitably be a commercially available 12 bit AIDconverter. The A/D converter 14 is further connected to the controller24 and the clock 26. The clock 26 provides the sampling frequency, whichis preferably approximately five times the IF carrier frequency.Accordingly, in an exemplary embodiment, the A/D converter 14 has asampling rate of 10 MHz.

In a preferred embodiment, a gate signal generator 28 controls or gatesthe A/D converter 14 such that the A/D converter 14 only digitizes aselect portion of the down-converted television signal. The selectportion of the television signal is preferably a portion in which thebaseband signal or information signal has predictable and repeatablecontent. In the exemplary embodiment described herein, the selectportion of the television signal is the portion of a television signalframe in which one of the vertical synchronization intervals is present.

In particular, the vertical synchronization interval is a portion of anNTSC standard television signal that has a predictable signal patternwhich is repeated twice per frame in the television signal, or sixtytimes per second. All of the vertical synchronization intervals of atelevision signal have substantially the same signal pattern, and occurat the same point of each video frame within the signal. In other words,the vertical synchronization signal occurs at regularly occurringintervals. In the present embodiment, the select portion of thetelevision signal corresponds to either, but not both, of the verticalsynchronization intervals in each television signal frame. The use ofonly one of the vertical synchronization intervals allows twice as muchtime for processing the select portion of the signal as would beavailable if both vertical synchronization intervals were used. Those ofordinary skill in the art may readily employ faster processing devicesor other modifications that would make the use of both verticalsynchronization intervals advantageous.

In any event, in the exemplary embodiment described herein, the gatesignal generator 28 provides a signal the enables the A/D converter 14only when the A/D converter 14 receives the portion of thedown-converted television signal that corresponds to a verticalsynchronization interval. Otherwise, the A/D converter 14 is disabled.

The A/D converter 14 digitizes the select portion of the down-convertedtelevision as described above to produce a plurality of digital signalsegments. Each digital signal segment has a length or duration definedby the length or duration of the select portion of the television signal(per frame). Accordingly, in the exemplary embodiment described herein,each digital signal segment has a length or duration that corresponds toa vertical synchronization interval of one standard NTSC televisionsignal frame. For reasons that will be discussed further below, thedigital signal segment should be selected such that it has a durationthat is equivalent to the duration of a plurality of horizontal lines ofa standard NTSC television signal.

The gating signal generator 28 provides the above described functions inthe following manner. As an initial matter, the gating signal generator28 is connected to receive the down-converted television signal from theanalog circuit 12, and is connected to provide an enabling signal orgating signal to an enable pin of the A/D converter 14. The gatingsignal generator 28 is further connected to the clock 26. In operation,the gating signal generator 28 first receives the down-convertedtelevision signal and demodulates the signal using a simple,inexpensive, and relatively imprecise analog demodulation technique,such as envelope detection. The gating signal generator 28 then filtersthe roughly demodulated television signal using a low pass filter havinga cutoff frequency sufficient to remove all content except the verticalsynchronization interval. For example, the gating signal generator 28may use a low pass filter having a cut off frequency of 10 kHz. Thegating signal generator 28 then uses any suitable comparator orthresholding circuit to detect the vertical synchronization interval.The gating signal generator 28 then determines, using the pulses fromthe clock 26, both the duration of the vertical synchronizationinterval, and the start time of each vertical synchronization interval(in terms of the clock pulses). The gating signal generator 28 uses suchinformation to generate a gating signal that is provided to the A/Dconverter 14.

The A/D converter 14 samples the down-converted television signal basedon the gating signal to generate one digital signal segment pertelevision frame. Each digital signal segment comprises a carriercomponent CARRIER modulated by an information signal component VIDEO,and further comprises a noise signal component NOISE. The signal CARRIERcorresponds to the IF carrier in the television signal, the signal VIDEOcorresponds to the information signal, for example, the verticalsynchronization interval, of the television signal, and the signal NOISEcorresponds to the noise present on the television signal.

The demodulator 18 receives the digital signal segment from the A/Dconverter 14 and performs digital demodulation thereon to separate thecarrier component from the information signal component and the noisesignal component. The digital demodulation techniques may suitably be acarrier recovery demodulation technique or a square law demodulationtechnique, both of which are discussed further below in connection withFIGS. 2 and 3, respectively. The combination of the information signalcomponent and the noise signal component that is left after demodulationis referred to herein as the baseband signal.

The demodulator 18 is operably connected to provide the basebandcomponent to the information signal extractor 20. The information signalextractor 20 performs digital processing techniques to remove theinformation signal from the baseband signal. To this end, theinformation signal extractor 20 generates or obtains an estimate of theinformation signal and essentially subtracts the information signalestimate from the baseband signal to produce the noise signal estimate.FIGS. 5A and 5B describe in further detail two exemplary embodiments ofthe information signal extractor 20.

The information signal extractor 20 provides the noise signal estimateto the noise processor 22. The information signal extractor 20 alsoprovides the information signal estimate to the noise calculator 23. Aswill be discussed further below, the noise calculator 23 uses theinformation signal estimate to generate a carrier level measurement,which is then used for carrier to noise measurements.

The noise processor 22 and noise calculator 23 in combination operate toreceive the noise signal estimate and perform digital signal processingon the noise signal estimate to provide any one of a plurality of selectnoise measurement operations. Specifically, a CATV noise signal mayinclude components such as hum noise, CTB noise, CSO noise, and snownoise. Accordingly, the noise signal estimate generated by theinformation signal extractor 20 includes a plurality of noisecomponents. The combination of the noise processor 22 and the noisecalculator 23 use digital signal processing techniques to substantiallyisolate one or more noise components for measurement of such componentsapart from other types of noise that may be present in the noise signalestimate.

For example, to obtain a CTB noise measurement from a noise signalestimate known to have CTB noise and hum noise, the noise processor 22may substantially reduce the portion of the noise signal energy due tohum noise. To this end, the noise processor 22 may employ digitalfiltering techniques to filter the noise signal estimate tosubstantially attenuate the frequencies at which hum noise is located.Such filtering may also be used to isolate CSO noise or snow noise. Oncethe select type of noise signal is isolated through filtering, the noisecalculator 23 measures the remaining noise signal energy using anysuitable energy measurement technique.

In a preferred embodiment, however, the noise processor 22 facilitatesthe measurement of a specific type of noise by first generating afrequency response of the noise signal estimate. The noise calculator 23then performs a noise energy calculation using only the portions of thegenerated frequency response that correspond to the types of noise beingmeasured. For example, if CTB noise is being measured, the noiseprocessor 22 first generates the frequency response of the noise signalestimate and the noise calculator 23 performs the noise energycalculation using only the portions of the frequency response between DCand 100 kHz.

To generate a frequency response, the noise processor 22 performs adiscrete Fourier transform (“DFT”) of the noise signal estimate. The DFTgenerates a frequency response comprising a plurality of frequency bins,and is defined by a resolution and a frequency range.

As is well known in the art, the frequency bins comprise scalarquantities representative of the amount of certain frequency componentspresent in the analyzed signal. For example, FIG. 7 shows a partialfrequency response of an exemplary noise signal estimate generated by anDFT. The partial frequency response of FIG. 7 includes a plurality offrequency bins including bin 0, bin 1, bin 2, bin 3 and bin 4. Eachfrequency bin provides an approximate level of the signal energy of thenoise signal estimate that is located within a particular frequencyband.

The resolution of the DFT, as is known in the art, defines the width ofthe frequency band represented by each bin, as well as the spacingbetween adjacent bins. The resolution of the DFT depends upon theduration of the analyzed signal segment. In the present embodiment, theduration of the analyzed signal segment is substantially equal to theduration of the vertical synchronization interval of a standard NTSCsignal. The vertical synchronization interval has a duration that isequivalent to nine horizontal lines of a standard television signal.Because horizontal lines occur at a frequency of 15.734 kHz, theduration of the vertical synchronization interval is (1/15734)*(9) or572 microseconds. Accordingly, in the present embodiment, the resolutionof the DFT is equal to 1/(duration), which would be 1/(572 u-sec) orapproximately 1.75 kHz.

As a result, each bin corresponds to a frequency band of 1.75 kHz, andare spaced 1.75 kHz apart, starting at DC. For example, as illustratedin FIG. 7, the first frequency bin, bin 0, is centered at DC andcorresponds to a band extending from −875 Hz to 875 Hz, or effectively,DC to 875 Hz. Bin 0 thus provides an estimate of the energy of theexemplary noise signal estimate that is located between DC and 875 Hz.The second frequency bin, bin 1, is centered at 1.75 kHz and extendsfrom 875 Hz to 2.725 kHz. Analogous to bin 0, bin 1 provides an estimateof the energy of the exemplary noise signal estimate that is between 875Hz and 2.725 kHz.

The frequency range of the frequency response defines the upperfrequency limit, in other words, the highest frequency for which afrequency bin is generated. As is known in the art, the frequency rangeof the DFT frequency response corresponds to the sampling rate of theanalyzed signal. In particular, the frequency range begins at DC andextends to one-half of the sampling rate. Thus, according to the presentembodiment in which the sampling rate is 10 MHz, the frequency range ofthe DFT generated by the noise processor is from DC to 5 MHz.

It will be noted that in order to further refine the noise measurementsgenerated by the present invention, it would be advantageous to increasethe duration of the select portion of the television signal. Theincreased duration would result in a finer resolution, which in turnincreases the precision of the noise measurement. To increase theduration of the select portion of the television signal, the selectportion could be defined to include both the vertical synchronizationinterval and one or more adjacent quiet lines.

Specifically, in NTSC standard television signals, the verticalsynchronization signal is typically followed by several quiet lines inwhich video information is not transmitted. Such quiet lines aretheoretically predictable and repeatable, similar in that respect to thevertical synchronization interval itself. If the select portion of thetelevision signal is defined as the vertical synchronization intervalplus one succeeding quiet line, the duration of the analyzed signalwould increase from nine horizontal lines to ten horizontal lines, andthe resolution would become (15734/10) or 1.5734 kHz.

Such a select portion definition, however, may not provide reliableresults. In particular, the quiet lines of NTSC television signals areoften employed to transmit text data or other information, and thereforeare not guaranteed to be predictable or repeatable. If the quiet line inthe defined select portion includes unpredictable and changinginformation, it cannot be used for accurate noise measurements.

It will further be noted that it may be preferable in some circumstancesto utilize fast Fourier transform techniques to generate the DFTfrequency response. A fast Fourier transform is a variation of a DFTthat has reduced computational time. Those of ordinary skill in the artmay readily implement fast Fourier transform if deemed necessary intheir particular implementation.

In any event, referring again to FIG. 1, once the noise processor 22generates the frequency response, the noise calculator 23 may thenisolate the noise signal estimate energy caused by a particular type ofnoise. In particular, the noise calculator 23 obtains an adjustedfrequency response that consists of frequency bins corresponding to thefrequencies in which the particular type of noise is located. The noisecalculator 23 effectively ignores the other frequency bins.

For example, if CTB noise is to be measured, the noise calculator 23obtains an adjusted frequency response that comprises only thosefrequencies at or below 100 KHz. Moreover, the noise calculator 23 takesfurther steps to remove the energy in the frequency response of thenoise signal estimate that is due to hum noise. In particular, the noisecalculator 23 removes the frequency bin corresponding to DC, such as bin0 of FIG. 7, to remove any hum noise component from the CTB noisemeasurement. Specifically, hum noise energy is concentrated around lowfrequency multiples of 60 Hz. Accordingly, substantially all of thenoise energy attributable to hum noise is located in bin 0. The noisecalculator 23 therefore removes bin 0 from the frequency response andreplaces it with an estimate of the CTB noise energy at the frequencyassociated with bin 0, or in other words, at DC. It has been determinedthat the CTB noise energy located in bin 1, i.e. between 875 Hz and2.625 kHz, is substantially similar to the CTB noise energy located inbin 0. Accordingly, the noise calculator 23 of the present inventionuses the frequency bin 1 as the estimate of frequency bin 0.

In another example, if CSO noise is to be measured, then the noisecalculator 23 uses only those frequency bins from the frequency responsethat fall within 100 kHz of each of 0.75 MHz and 1.25 MHz.

In addition, if snow noise is to be measured, then the noise calculator23 ignores all the frequency bins from the frequency response that couldpossibly contain CSO, hum or CTB noise. Accordingly, the noisecalculator 23 ignores all frequency bins except those that correspond tofrequencies between 100 kHz and 700 kHz. It should be noted that becausesnow noise energy is relatively evenly distributed across the frequencyspectrum, sufficient information regarding snow noise may be determinefrom a 100 kHz frequency band located between 100 kHz and 700 kHz. As aresult, the noise calculator 23 may suitably determine the snow noiselevel using frequency bins that correspond to a 100 kHz (or evennarrower) frequency band located between 100 kHz and 700 kHz.

The noise calculator 23 then generates a noise power measurement for theparticular type of noise by integrating the adjusted frequency response.

In particular, to generate a measurement of CTB noise from a properlyadjusted frequency response, the noise calculator 23 takes integrals ofthe noise power through a 30 kHz window moving from DC to 100 kHz. Theintegral of the 30 kHz frequency band having the maximum noise is usedas the CTB noise power measurement.

In an example in which a measurement of CSO noise is to be generated,the noise calculator 23 takes integrals of the noise power through a 30kHz window moving from 0.7 MHz to 0.8 MHz and also moving from 1.2 MHzto 1.3 MHz. The integral of the 30 kHz band having the maximum noise isused the CSO noise power measurement.

The noise calculator 23 also determines the carrier power level usingthe information signal estimate received from the information signalextractor 20. To this end, the noise calculator may perform a root meansquares power calcuation or the like on the peaks of the receivedinformation signal estimate. The noise calculator 23 then provides tothe display 25 the ratio of the noise power measurement to carrier powerlevel.

Accordingly, the present invention provides a method of measuring noisein which different types of noise may be isolated and measured usingdigital signal processing techniques. By using DFT techniques togenerate a frequency response of the noise signal estimate energy, thenoise signal energy due to any of the particular types of noise may begenerated. Moreover, the frequency response may further manipulated asdescribed above to remove virtually all hum noise components from theCTB measurement, thereby increase measurement accuracy.

The use of the frequency response further facilitates the removal ofsnow noise power, which is present at all frequencies, from themeasurement of CSO and CTB noise. If removal of the the snow noise powerfrom a measurement of either CSO or CTB noise is desired, the snow noisepower may be measured and then subtracted from the CSO or CTB noisepower measurement generated in the manner described above.

For example, consider a first measurement generated in the mannerdescribed above in connection with the generation of a CTB noisemeasurement. Because snow noise is typically present at all frequencies,the first measurement represents the CTB noise measurement combined witha measurement of the snow noise that is found in the frequency range ofthe first measurement. To remove the portion of the first measurementthat is attributable to snow noise, the noise processor 22 provides anadjusted frequency spectrum to the noise calculator 23 that contains a100 kHz frequency band located between 100 kHz and 700 kHz, where snownoise alone is located. The noise calculator 23 then generates a snownoise measurement using the same integration techniques.

In particular, the noise calculator 23 obtains integrals of the noisepower through a 30 kHz window moving through the 100 kHz snow noisefrequency band. The average integral of the 30 kHz bands is then used asthe average snow noise measurement. The average snow noise measurement,which represents the snow noise over 30 kHz, may then be subtracted fromthe first measurement to remove the influence of the snow noisetherefrom. The resulting measurement represents an enhanced CTB noisemeasurement with little or no snow noise contributing to themeasurement.

It will be noted, however, that snow noise is often negligible andtherefore typically does not have to separated from the initial CTB orCSO noise power measurements using the methods described above.

If snow noise itself is to be measured, then the noise calculator 23extrapolates the average snow noise measurement over the entire spectrumbeing measured. In the present embodiment, the snow noise may bedetermined for a CATV channel, which may be considered to have a 4.2 MHzbandwidth. To obtain the snow noise for the entire band, the averagesnow noise measurement is multiplied by (bandwidth/30 kHz), or 140.

Regardless the type of noise being measured, the present invention thusprovides several advantages over prior in-service noise testing devices.In particular, by generating a frequency response of the noise signalestimate, the present invention provides a means by which certains typesof noise may be measured in an otherwise combined noise signal.Moreover, the use of digital processing methods to generate thefrequency response eliminates the need for analog filters to isolatevarious types of noise for measurement. The reduction of analog filtersreduces component costs for the device. Similarly, the use of digitalcarrier recovery techniques, instead of analog demodulation, eliminatesthe need for, and cost associated with, analog demodulation circuitry.

The above embodiment of the present invention further provides enhancedaccuracy of measurement by using the vertical synchronization intervalas the basis for the digital signal segment. In addition to being apredictable and repeating pattern, which facilitates isolation of thenoise signal in an in-service channel, the vertical synchronizationinterval is the longest contiguous predictable and repeating pattern inthe standard NTSC television signal frame. Such relative length of thevertical synchronization interval allows for the DFT-generated frequencyresponse to have a sufficiently fine resolution for accuratemeasurement.

In particular, the fine resolution offered by the present embodiment,1.75 kHz resolution, allows for the first bin (bin 0) to be removed fromthe frequency response and be replaced by an estimate thereof to reducethe effects of hum noise in the CTB measurement, as discussed above. Theremoval of bin 0 is possible because the remaining bins provide enoughinformation about the CTB noise spectrum to allow estimation of the CTBnoise that should be present in the frequencies represented by bin 0.If, by contrast, a digital signal segment corresponding to a singlehorizontal line is used, then only 15.734 kHz resolution is provided. Insuch a case, if bin 0 is removed to eliminate the effects of hum, theremaining bins do not provide sufficient information to estimate the CTBnoise. In particular, bin 0 in a frequency response having 15.734 kHzresolution will represent the noise signal in frequencies from DC toapproximately 7.67 kHz. In some cases, almost all of the CTB noise islocated in that frequency range, and thus the removal of bin 0 wouldprevent any measurement of CTB noise. As a result, the use of a digitalsignal segment consisting of a single horizontal line of a televisionsignal will not produce a sufficient frequency spectrum to allow theremoval of hum noise from the CTB noise measurement.

FIGS. 2, 3, 4A, 4B, 5A and 5B, discussed below, illustrate in furtherdetail of the functions performed by the DSP 16, as well as the flow ofinformation, in other words, digital signals, between the functionsperformed by the DSP 16. For purposes of clarity, operations of the DSP16 are represented by functional blocks. While the functional blocks aregiven device names, their function is typically carried out byprogramming the DSP 16 to perform the described function. It will beappreciated, however, that those of ordinary skill in the art mayreadily utilize discrete digital components to carry out a part or allof the functions associated with one or more of the functional blocksillustrated in FIGS. 2, 3, 4A, 4B, 5A and 5B.

FIG. 2 shows a first embodiment of the demodulator 18 of FIG. 1 Thedemodulator of FIG. 2 is a carrier recovery deodulator 18 that comprisesan input 102, a carrier recovery block 104, a multiplier 106, a low passfilter 108 and an output 110. As discussed above, the demodulator 18recieves the digital video signal, or digital signal segment, andproduces a baseband signal therefrom. As illustrated in FIG. 2, thedemodulator 18 receives the digital signal segment at the input 102 andprovides the baseband component, or baseband signal at its output 110.

The digital signal segment received at the input is fairly approximatedby the quantity:DSS=CARRIER*VIDEO+NOISE,where CARRIER is the down-converted carrier signal (at the intermediatefrequency), VIDEO is the information signal, and NOISE is the noisesignal, which may include CTB and CSO components. It is noted that thesignal CARRIER may further include, among other things, a hum noisecomponent. The signal provided at the output 110 will be a basebandsignal that includes ½(VIDEO+NOISE′), where NOISE′ is a frequencyconverted version of the signal NOISE, and may further includes the humnoise from the signal CARRIER.

In mathematical terms, the DSS may be approximated by the followingrelationship.DSS=TV(t)cos(wt)+a cos((w+w′)t)where TV(t) is VIDEO, cos(wt) is CARRIER, and a cos(w′t) is NOISE(including a scalar component a and a frequency component cos(w′t)). Itis noted that the above approximation is simplified for the purposes ofassisting in explaining the theory of operation of the presentinvention, and does not include all components of a true digital videosignal segment. For example, the signal NOISE is represented as having asingle frequency component, w+w′, and the phase components to CARRIERand NOISE are omitted. Such simplifications facilitate the theoreticalexplanation.

The carrier recovery block 104 is a functional block that generates areliable estimate CARRIER_EST of the carrier signal CARRIER. CARRIER_ESTin the above terms may be written as cos(w″t), where w″ is a frequencyvalue close to if not substantially identical to the value w. The stepsrequired to carry out the function of the carrier recovery block 104 arediscussed further below in connection with FIGS. 4A and 4B. In summary,however, the carrier recovery block 104 provides the estimateCARRIER_EST to a multiplier 106, while the signal DSS iscontemporaneously provided to the multiplier 106. The multiplier 106then multiplies DSS and CARRIER_EST to generate a product PROD:

PROD = CARRIER_EST * [(CARRIER * VIDEO) + NOISE] = cos(w”t) *{TV(t)cos(wt) + acos((w + w)’t)} = TV(t)cos(wt)cos(w”t) + acos((w +w’)t)cos(w”t) = (½)TV(t){cos[(w − w”)t] + cos[(w + w')t]} + (a/2){cos[(w− w” + w’)t] + cos[(w + w” + w’)t]} = (½){TV(t)cos[(w − w”)t] + acos[(w− w” +w’)t]} + (½){TV(t)cos[(w + w”)t] + acos [(w + w” + w’)t]}Because the carrier recovery block 104 generates a fairly accurateestimate of the signal CARRIER, or in other words, w″=w, then, the valuew may be substituted for w″ and the following equation results:PROD=(½){TV(t)+a cos(w′t)}+(½){TV(t)cos(2wt)+a cos [(2wt+w′)t]}Thus, PROD=(½)(VIDEO+NOISE′)+(½)(CARRIER″*VIDEO+NOISE″), where CARRIER″is a signal having approximately twice the frequency of the signalCARRIER, NOISE′ is a down-converted version of the signal NOISE, andNOISE″ is a version of the noise signal having twice the frequency ofthe signal CARRIER.

The signal PROD is then provided to the low pass filter 108, which has acut-off frequency below the carrier frequency, thereby providing thesignal BASEBAND, where BASEBAND=½(VIDEO+NOISE′). The signal BASEBAND isthen provided to the output 110, which may subsequently be provided tothe information signal extractor 20 (see FIG. 1).

FIGS. 4A and 4B show alternative embodiments of the carrier recoveryblock 104 that is used in the carrier recovery demodulator of FIG. 2.FIG. 4A shows a first embodiment of a carrier recovery block 104 thatuses a zero crossing estimator to obtain the frequency of the signalCARRIER. FIG. 4B shows a second embodiment of a carrier recovery block104 that uses a rough estimate of the signal CARRIER and then refinesthe rough estimate.

Referring to FIG. 4A, the carrier recovery block 104 includes an input202, an output 204, a zero crossing estimator 206, a continuous wave(“CW”) generator 208, first and second multipliers 210 and 212,respectively, first and second low pass filters 216 and 218,respectively, a phase isolator 220, and a second CW generator 222.

The input 202 is connected to each of the first multiplier 210, the zerocrossing estimator 206, and the second multiplier 212. The first andsecond multipliers 210 and 212, respectively, each are operable tomultiply two digital input signals together to generate an outputsignal. The zero crossing estimator 206 is operable to receive a digitalsignal and generate an estimate of the frequency of the received digitalsignal by detecting and analyzing the zero crossings therein. Those ofordinary skill in the art may readily employ digital signal processingalgorithms to carry out the functions described above in connection withthe zero crossing estimator 206 and the multipliers 210 and 212.

The zero crossing estimator 206 further connected to both the first andsecond CW generators 208 and 222, respectively. The first and second CWgenerators 208 and 222 are operable to receive as inputs both frequencyand phase information and produce therefrom a sine and/or cosine wavesignal having a frequency and phase corresponding to the frequency andphase information received at their inputs.

An exemplary CW generator may be implemented in a DSP-basedconfiguration in the following manner. Generally, the CW generator usesa look-up table, stored within a memory, that contains a digitalrepresentation of a sine wave. For example, the look-up table mayinclude 360 entries, each entry representing the instantaneous magnitudevalue of a sine wave for a particular degree in a cycle. The CWgenerator then receives as inputs, frequency and phase information, aswell as a clock signal. The CW generator then uses the frequencyinformation to determine how fast to progress through the table. Morespecifically, the frequency information multiplied by the clockinformation, added to the phase information, identifies exactly whichvalue of the look-up table should be produced at the output of the CWgenerator. Those of ordinary skill in the art may readily program adigital signal processing device or otherwise configure discreteelements to perform the functions described above in connection with theCW generators 208 and 222.

The first CW generator 208 is further connected to the first multiplier210 and the second multiplier 212. The output of the first multiplier210 is provided to the first low pass filter 216 and the output of thesecond multiplier 212 is provided to the second low pass filter 218. Thefirst and second low pass filters 216 and 218, respectively, are eachfurther connected to the phase isolator 220. The first and second lowpass filters 216 and 218, respectively, may suitably be ordinary digitallow pass filters, such as finite or infinite impulse response filters.The first and second low pass filters 216 and 218 each have a cut offfrequency below twice the frequency of the signal CARRIER. The first andsecond low pass filters 216 and 218 should each have at least 70 dB ofattenuation in the stop band.

The phase isolator 220 is a device that receives a time-varying phasesignal, such as sin [p(t)], and a 90° phase-shifter version of thatsignal cos [p(t)] and produces the time-varying phase angle or value,p(t), as an output. The phase isolator 220 may readily be implementedusing digital signal processing techniques. In the present embodiment,the phase isolator 220 receives both a sin [p(t)] value and a cos [p(t)]value corresponding to each digital sample corresponding to or indexedby t. For each digital sample, the sign of each of those signals isfirst examined to determine the quadrant in which time-varying phasecomponent may be found. Then, the first signal is divided by the secondsignal, which produces a tangent value of the time-varying phasecomponent. Then, the inverse tangent of the tangent value is taken,which, combined with the quadrant information, determines the preciseinstantaneous phase angle for that sample. In summary, the phaseisolator 220 receives a signal sin [p(t)] at one input and cos [p(t)] atthe other input and generates a value representative of the value of pfor each sample indexed by t.

For example, consider an example wherein the instantaneous value ofp(t)=150°. The phase isolator 220 receives at its two inputs sin(150°)and cos(150°), or 0.5 and −0.87, respectively. The phase isolator 220first determines that because the sine value is positive and the cosinevalue is negative, the angle p(t) is in the second quadrant. The phaseisolator 220 then divides 0.5 by −0.87 and takes the arctangent of theresult, yielding p(t)=150°.

The operation of the carrier recovery block 104 illustrated in FIG. 4Ais described herebelow. As an initial matter, it is noted that the inputsignal is CARRIER*VIDEO+NOISE, as discussed above in connection withFIG. 2. For the purposes of discussing the removal of the carrier inconnection with FIG. 4A, (CARRIER*VIDEO)+NOISE may be written asV(t)cos(wt+p), where V(t) is equal to the information signal, VIDEO, andcos(wt+p) is the signal CARRIER. The NOISE signal is sufficiently smallcompared to the signal VIDEO such that it may be ignored during thediscussion of the carrier recovery block 104. It is noted that the inputsignal is represented in a different form for this discussion of FIG. 4Athan it was for the discussion of the theory behind the circuit in FIG.2. Both equations are essentially accurate, except that FIG. 2 requiredthat the VIDEO and NOISE signals be represented separately, and thephase of the signal CARRIER was not important. By contrast, thedistinction between the VIDEO and NOISE signals is not relevant to thediscussion of FIG. 4A, but the phase of the signal CARRIER is important.

In operation, the input 202 provides V(t)cos(wt+p) to each of the firstmultiplier 210, the second multiplier 212, and the zero crossingestimator 206. The zero crossing estimator 206 determines the frequencyof the signal, w, using analysis of the zero crossings of V(t)cos(wt+p).The zero crossing estimator 206 generates a digital value representativeof w and provides w to both the first and second CW generators 208 and222, respectively. The first CW generator 208 generates both a sine waveand a cosine wave using the frequency value w received from the zerocrossing estimator 206. No phase information is provided to the first CWgenerator 208, and thus the fist CW generator 208 provide output signalswith no phase component. Specifically, the outputs are sin(wt), which isprovided to the first multiplier 210, and cos(wt), which is provided tothe second multiplier 212.

The first multiplier 210 thus receives V(t)cos(wt+p) and sin(wt) andmultiplies those two signals together. The result of the multiplicationis ½V(t){ sin(2wt+p)−sin p}. The low pass filter 216 removes all but thecomponent −½V(t)sin p, which is then provided to the phase isolator 220.Similarly, the second multiplier receives V(t)cos(wt+p) and cos(wt) andmultiplies those signals together. The result of the multiplication is½V(t){ cos(2wt+p)+cos p}. The second low pass filter 218 removes all bythe component ½V(t)cos p, which is then also provided to the phaseisolator 220. The phase isolator 220 operates as discussed above togenerate the phase angle, p, from the received signals ½V(t)cos p and−½V(t)sin p. The phase isolator 220 provides the phase angle p to thesecond CW generator 222. The second CW generator 222 uses the receivedphase angle p and the value w received from the zero crossing estimator206 to generate a signal cos(wt+p). The signal cos(wt+p) represents thesignal CARRIER_EST. Although the output of the second CW generator 222is shown to be substantially equivalent to the signal CARRIER, suchoutput will actually only be the estimate, CARRIER_EST. The estimatesignal, CARRIER_EST, may then be utilized as described above inconnection with FIG. 2.

FIG. 4B shows an alternative embodiment of a carrier recovery block 254,which may suitably be substituted for the carrier recovery block 104 inFIG. 2. The alternative carrier recovery block 254 has a substantiallysimilar configuration as the carrier recovery block 104 of FIG. 4A witha few exceptions. For ease of reference, identical components in FIGS.4A and 4B are denoted with identical reference numerals. The onlydifferences between the carrier recovery block 104 and the alternativecarrier recovery block 254 are that the zero crossing estimator 206 ofFIG. 4A is replaced with a carrier estimator 260, and the input 202 isnot connected to the carrier estimator 260.

In operation of the alternative carrier recovery block 254, the input202 receives the signal V(t)cos(wt+p), as above, and provides the signalto the first and second multipliers, 210 and 212, respectively. Thecarrier estimator 260 provides an estimate, x, of the carrier frequency,w, to the first CW generator 208 and the second CW generator 222. Tothis end, because the carrier frequency does not change, and thus theestimate, x, does not change, the carrier estimator 260 may simply be apermanent stored value equal to the carrier frequency estimate. In anyevent, the first CW generator 208 provides two outputs using thefrequency estimate, x. Specifically, the first CW generator 208 providesa sine wave signal, sin(xt) to the first multiplier 210, and a cosinewave signal cos(xt) to the second multiplier 212.

The first multiplier 210 multiplies the signals V(t)cos(wt+p) andsin(xt), thereby producing the signal ½V(t){ sin [(w+x)t+p]−sin[(w−x)t+p]}. The first low pass filter 216 substantially removes allcomponents except for the signal −½V(t)sin [(w−x)t+p], which is thenprovided to the phase isolator 220. Similarly, the second multiplier 212multiplies the signal V(t)cos(wt+p) with cos(xt), thereby producing thesignal ½V(t){ cos [(w+x)t+p]+cos [(w−x)t+p]}. The second low pass filter218 substantially removes all components except for the signal ½V(t)cos[(w−x)t+p], which is then also provided to the phase isolator 220. Thephase isolator 220 receives the signals −½V(t)sin [(w−x)t+p] and½V(t)cos [(w−x)t+p], and produces the value (w−x)t+p, which is thenprovided to the second CW generator 222.

The second CW generator 222 receives the phase value (w−x)t+p and theestimated frequency x from the carrier estimator 260 and produces thesignal cos(wt+p) therefrom. The signal cos(wt+p) is the signal CARRIER,or more accurately, an estimate, CARRIER_EST, of the signal CARRIER. Thesignal CARRIER_EST may then be used as described above in connectionwith FIG. 2.

The carrier recovery blocks 104 and 254 illustrated in FIGS. 4A and 4B,respectively, each provide a highly accurate estimate of the carriersignal which allows the received signal to be sampled at a relativelylow sampling rate such as that specified for the A/D converter 14 ofFIG. 1. The sampling rate, as discussed above in connection with FIG. 1,may suitably be on the order of five times the IF carrier frequency.Such a low sampling rate is not compatible with other digitaldemodulation techniques, such as digital envelope detection. In envelopedetection, the peaks of the digitally sampled signal are analyzed andpassed through a low pass filter in order to demodulate the signal. Ifonly five samples are provided for every cycle of the signal, then incannot be guaranteed that the peak value of the signal will bedigitized. As a result, such a low sampling rate can introducedistortion in an envelope detection demodulation process. Suchdistortion may add substantially to the measured noise component of thesignal and in fact effectively destroy the integrity of the measurement.

Nevertheless, it will be appreciated that the use of a higher samplingrate to IF carrier frequency ratio would facilitate the use of anenvelope detector circuit instead of the carrier recovery demodulatorillustrated in FIG. 2. However, increasing that ratio requiresincreasing the sampling rate, which can add to the cost ofimplementation.

In any event, the use of any digital demodulation technique reduces thecost normally associated with analog demodulation circuitry. Inparticular, analog demodulation circuitry may be any number of devices,including a log amp detector. The digital demodulation techniquesdescribed above utilize digital signal processing circuitry, such as adigital signal processor, that is already included in the circuit forother reasons. Thus, the use of digital demodulation can eliminate therequirement for special analog demodulation circuitry without requiringadditional digital circuit components.

FIG. 5A shows an information signal extractor 20 for use in connectionwith a first embodiment of the present invention. In particular, theinformation signal extractor 20 of FIG. 5A is used in the noisemeasurement circuit 10 of FIG. 1. The information signal extractor 20 ofFIG. 5A is compatible for use with, among other things, the embodimentof the demodulator 18 described above in connection with FIG. 2. Theinformation signal extractor 20 includes an input 301, a synchronizer302, an information signal estimator 304, an adder 306, and an output308.

In general, the information signal extractor 20 receives the basebandsignal BASEBAND. The baseband signal includes an information signal,VIDEO, and a noise signal, NOISE′. Specifically, as discussed above inconnection with FIG. 2, BASEBAND=½(VIDEO+NOISE′) in the presentembodiment, where NOISE′ is a frequency converted version of theoriginal signal NOISE. The signal NOISE′ is down-converted by afrequency equivalent to the IF carrier frequency, which in the presentembodiment is 2 MHz. The information signal VIDEO comprises primarilythe vertical synchronization interval of a standard television signal.In any event, the information signal extractor 20 produces at its output308 a signal consisting of an estimate of the noise signal, NOISE′.

To this end, the signal BASEBAND is first provided by the input 301. Theinput 301 provides the signal BASEBAND to each of the information signalestimator 304 and the adder 305 through a synchronizer 302. Thesynchronizer 302 operates to align the information signal within thedigital signal segment such that the start of the verticalsynchronization interval coincides with a particular sample in thedigital signal segment.

In particular, to carry out the functions described below, theinformation signal extractor 20 requires that the start of the verticalsynchronization interval occur at a specified sample, for example, thetenth sample of the digital signal segment. However, the gating signalgenerator 28 of FIG. 1 is in practicality not capable of insuring thatthe vertical synchronization interval starts precisely at the samesample in every digital signal segment. In fact, the start verticalsynchronization interval in the present embodiment may vary up to 10microseconds within the digital signal segment, or 100 samples.Accordingly, the synchronizer 302 is employed to provide sufficientdelay to ensure that the vertical synchronization interval always startson the same sample within the digital signal segment. To this end, thesynchronizer 302 may suitably be a match filter that performs acorrelation with an ideal vertical synchronization interval to detectthe beginning of the vertical synchronization interval in the inputsignal BASEBAND. Those of ordinary skill in the art may readilyimplement such a match filter. It shall be noted that the synchronizer302 does not affect that spectral qualities of the signal BASEBAND inany way.

In any event, the synchronizer 302 provides the time synchronized signalBASEBAND to the information signal estimator 304 and the adder 305. Theinformation signal estimator 304 then generates an estimate of thesignal VIDEO using an average of a plurality of previous digital signalsequences, for example, the last N digital signal sequences. To thisend, the information signal estimator 304 stores and averages a numberof input digital signal sequences, ½(VIDEO+NOISE′), over time. Eachsequence comprises a plurality of samples that preferably correspond tothe entire digital signal segment. As discussed above in connection withthe operation of the A/D converter 14 of FIG. 1, the digital signalsegment is selected such that the information signal is a predictablerepetitive waveform, such as the vertical synchronization interval of anNTSC video signal. The use of a repetitive waveform facilitates theestimation of the information signal. Because the content of the signalVIDEO is a known and repeating waveform, the VIDEO term should always bethe same value. It has been observed that the average of the NOISE′component over several sequences is substantially zero, or at leastsufficiently small enough that it does not compromise the accuracy ofmeasurements. Thus the average of the plurality of digital signalsequences is equal to the VIDEO term without the NOISE term. Theinformation signal estimator 304 then provides the average quantity, andmore specifically, the inverted average quantity, −½VIDEO, to the adder305.

The adder 305 thus receives the combined noise and video basebandsignal, BASEBAND, and the inverted average quantity −½VIDEO, and addsthe signals, thereby producing a resultant signal ½NOISE′. The adder 305provides the signal ½NOISE′ to the output 308. The output 308 providesthe signal ½NOISE′ to a noise processor, such as the noise processor 22of FIG. 1. The signal ½NOISE′ may suitable be adjusted by a scalarquantity either in the noise processor 22 or otherwise to obtain thevalue NOISE′.

It will be noted that instead of using the vertical synchronizationinterval portion of the video signal, a quiet line of the NTSC videosignal may be used. In each NTSC video signal frame, there are aplurality of quiet lines that consist of a DC offset and a horizontalsynchronization pulse. To use the quiet lines, the controller 24 of FIG.1 would be configured to cause the A/D converter of FIG. 1 to digitizeportions of the down-converted television signal that contain the quietlines. In such a system, the signal BASEBAND would be substantiallycomposes of ½NOISE′ with a DC offset and the horizontal synchronizationpulse. However, use of the quiet lines has some drawbacks. One drawbackis that different systems use some or all of the potential quiet linesto transmit other information. Accordingly, a system that uses quietlines cannot predict which potential quiet lines will actually be quietlines in a particular implementation.

Moreover, in order to achieve the measurement accuracy of the systemthat uses the vertical synchronization interval, several quiet lineswould have to be used. In particular, the measurement accuracy of thenoise measurement system 10 of FIG. 1 depends in part on the number ofsamples in the digital signal segment. The digital signal processingtechniques to distinguish the several types of noise, for example, fastFourier transforms, provide results having an accuracy that increases asa function of the number of samples taken. Because the verticalsynchronization interval is the equivalent of nine horizontal lineslong, nine quiet lines would be required to achieve results of similaraccuracy using quiet lines. Because the quiet lines are often used forother purposes, nine quiet lines per video frame are not reliablyavailable. As a result, the use of the vertical synchronization intervalis preferred over the use of quiet lines.

However, in one embodiment using quiet lines, several quiet lines ofeach NTS video frame may be sampled by the A/D converter 14,demodulated, and provided to the information signal extractor 20. Insuch an embodiment, the same method for isolating the signal NOISE′ asdescribed above may be used. In other words, the information signalestimator 304 would receive and average N groups of M quiet lines from Nvideo frames to produce an estimate of the information signal. Asdiscussed above, taking the average effectively removes the noise andgenerates a reliable estimate of the information signal. The signalestimate (scaled as necessary) may then be provided to the adder 305 tobe subtracted from the signal BASEBAND.

In any event, the embodiment of FIGS. 1, 2 and 5A provides a method ofdigitally demodulating a digital signal segment that includes a carriercomponent modulated by an information signal plus noise,(CARRIER*VIDEO)+NOISE, wherein the demodulated digital signal segment,or baseband signal BASEBAND, includes an information component (VIDEO)and a noise component (NOISE′). That embodiment further removes theinformation signal component from the demodulated baseband signal, inorder to generate a noise signal estimate. As discussed above, digitalsignal processing methods may then be used to measure components of thenoise signal estimate (NOISE′) that are due to CTB noise, CSO noise,and/or snow noise.

FIGS. 3 and 5B show alternative embodiment of the preset invention, andin particular, show alternative configurations for demodulator 18 andthe information signal extractor for use in the noise measurementcircuit 10 of FIG. 1. FIG. 3 shows a square demodulator 150 which mayreadily be employed as the demodulator 18 in the circuit of FIG. 1. Thesquare law demodulator 150 may therefore be employed as an alternativeto the carrier recovery demodulator 18 described above in connectionwith FIG. 2. The square law demodulator 150 includes an input 152, amultiplier 154, a low pass filter 156, and an output 158. The low passfilter 156 has a stop band defined as extending upwards from the IFcarrier frequency, and preferably has at least 70 dB of attenuation inthat defined stop band.

In general, the input 152 receives the digital signal segment DSS,which, as discussed above, is equal to (CARRIER*VIDEO)+NOISE. The signalDSS is then provided to both inputs of the multiplier 154. Because thesignal DSS is provided to both inputs to the multiplier 154, themultiplier effectively squares the signal DSS. The squared DSS signalproduces high frequency components and a baseband component. The lowpass filter 158 receives the squared DSS signal and substantiallyremoves all components except for the baseband component, which is(½)VIDEO*(VIDEO+2(NOISE′)).

In particular, if the DSS signal is represented as discussed above inconnection with FIG. 2, then

DSS = TV(t)cos(wt) + acos[(w + w’)t] where TV(t) is VIDEO, cos(wt) isCARRIER, and acos[(w +w’)t] is NOISE. The multiplier 154 multipliesDSS * DSS, which produces: DSS² ={TV(t)cos(wt) + acos[(w + w’)t]}² =TV²(t)cos²(wt) + 2TV(t)acos(wt)cos[(w +w’)t] + a²cos²[(w + w’)t]=(TV²(t)/2) {1 + cos(2wt)} + TV(t)a{cos(w’t) + cos[(2w +w’)t]} +(a²/2){1 + cos[2(w + w’)t]} =(½){TV²(t) + 2TV(t)acos(w’t) + a² } +(½){TV²(t)cos(2wt) + 2TV(t)acos[(2w + w’)t] + a²cos[2(w + w’)t]}

The above signal is then provided to the low pass filter 156, whichremoves the high frequency component and leaves (½){TV²(t)+2TV(t)acos(w′t)+a²}. The scalar of the noise signal, a, is generally so muchsmaller than TV(t) that the a² term drops out of the above equation,leaving (½({TV²(t)+2TV(t)a cos(w′t)}, or TV(t)/2*{TV(t)+2 a cos(w′t)},which may be written as (½)VIDEO*(VIDEO+2(NOISE′)). As before NOISE′ isa down-converted version of the original signal NOISE, plus any humnoise from the signal CARRIER.

The signal (½)VIDEO*(VIDEO+2(NOISE′)) is then provided to the output158. Although the signal (½)VIDEO*(VIDEO+2(NOISE′)) is not equivalent inform to the signal BASEBAND discussed above in connection with FIG. 2.Nevertheless, the baseband signal in this embodiment, similar to thatdescribed above in connection with FIG. 2, contains both an informationsignal component (VIDEO), a noise signal component (NOISE′), and nocarrier signal. This baseband signal is then provided to an informationsignal remover such as the one illustrated in FIG. 5B.

Referring to FIG. 5B, an information signal remover 350 includes aninput 351, a synchronization filter 352, an information signal estimator354, a divider 356, an adder 358 and an output 360. In operation, theinformation signal remover receives the baseband signal in the form of(½)VIDEO*(VIDEO+2(NOISE′)) at the input 352 and generates the signalNOISE′ at the output 360.

Specifically, the input 351 provides the input signal(½)VIDEO*(VIDEO+2(NOISE′)) to each of the divider 356 and theinformation signal estimator 354 through the synchronization filter 352.The synchronization filter 352 operates as described above to ensurethat the start of the information signal (squared), VIDEO² occurs at aparticular sample within the baseband signal.

The information signal estimator 354 operates in the same general manneras the video signal estimator 304 of FIG. 5A, discussed above.Specifically, the information signal estimator 354 stores and averagesseveral signal sequences that correspond to known repetitive signalsequences of the underlying video signal, such as the verticalsynchronization interval. In contrast to the information signalestimator 304 of FIG. 5A, however, the information signal estimator 354of FIG. 5B averages the signal (½)VIDEO*(VIDEO+2(NOISE′)), whichprovides an average of the signal (½)VIDEO² +(VIDEO*NOISE′). As before,the random nature of the signal NOISE′causes the term VIDEO*NOISE′ todrop out in an average, leaving on the signal (½)VIDEO². Thus, toprovide a true estimate of the signal VIDEO, the information signalestimator 354 further performs a square root operation on the averagesignal (without the ½scalar) quantity, VIDEO², producing an estimate ofthe signal, inverts the estimate signal, and provides the resulting,−(½)VIDEO to the adder 358.

The divider 356 thus receives the input signal(½)VIDEO*(VIDEO+2(NOISE′)) and divides that input signal by the signalVIDEO received from the information signal estimator 354. The divider356 provides the resulting signal, (½)VIDEO+NOISE′, to the adder 358.The adder 358 add the signal (½)VIDEO+NOISE′ to the signal −(½)VIDEOreceived from the information signal estimator 354 to produce theresulting signal, NOISE′. The adder 358 then provides the signal NOISE′to output 360. The output 360 may then provide the signal to a noiseprocessor such as the noise processor 22 discussed above in connectionwith FIG. 1.

The alternative embodiment described in connection with FIGS. 3 and 5Babove provide an advantage over the first embodiment because the squarelaw demodulator 150 of FIG. 3 does not require recovery of the carriersignal, which is a computationally complicated process. The drawback,however, is that the alternative embodiment adds some complexity to theinformation removal stage because square root and division operationsare involved. Depending on the particular hardware used, eitherembodiment may be more efficient (time or cost) over the other.

It will be understood that the above-described embodiments are given byway of example only, and that those of ordinary skill in the art mayreadily devise their own implementation that incorporate the principlesof the present invention and fall within the spirit and scope thereof.For example, those of ordinary skill in the art may readily employanalog demodulation techniques to remove the carrier signal from thebaseband signal. Selected portions of the baseband signal may then bedigitized and provided to the information signal extractor 20 and noiseprocessor 22. Suitable analog demodulation techniques typically requirerectification of the analog signal, and such rectification addsnonlinear distortion to the signal.

Furthermore, the information signal estimator 304 of FIG. 5A may bereplaced by an information signal estimator that does not average aplurality of digital signal segments, but rather stores an estimate of aknown information signal component of a digital signal segment. Forexample, if the digital signal segment is chosen to correspond to thevertical synchronization interval of the video signal, the informationsignal estimator may store an estimate of a generic verticalsynchronization interval sequence and provide that stored estimate tothe adder 306. It is known, however, that the vertical synchronizationinterval varies to some degree from signal to signal. Specifically, thepulse width of the various pulses in the vertical synchronizationinterval can vary from channel to channel. The alternative informationsignal estimator would be able to determine the variations from thesignal BASEBAND and adjust the estimate accordingly before providing theestimate to the adder 306. For example. While the general knowledge ofthe vertical synchronization interval is known, a comparator orthresholding technique may be used to gain specific knowledge of thepulse width variation with a particular vertical synchronizationinterval sequence. Those of ordinary skill in the art may readilygenerate a suitable information signal estimate using a predeterminedestimate of the vertical synchronization interval and the signalBASEBAND.

In addition, the above configuration describing the operation of the DSP16 are given by way of example only. Those of ordinary skill in the artmay readily implement alternative configurations to carry out the samefunctions as those described above.

1. A method of determining noise in a CATV channel, the CATV channelcomprising a predetermined frequency band, the CATV channel beingin-service when a transmitted carrier signal having a frequency in thepredetermined frequency band is present, the method comprising: a)obtaining a television signal corresponding to a predetermined CATVchannel, the television channel comprising a carrier signal modulated byan information signal; b) sampling at least a part of the televisionsignal to produce a digital signal segment, said digital signal segmentcomprising a noise signal component and an information signal component,wherein said information signal component has a substantiallypredetermined signal pattern; c) processing the digital signal segmentto produce a baseband signal; and d) obtaining an estimate of theinformation signal and subtracting the estimate from the basebandsignal, thereby producing a noise signal estimate.
 2. The method ofclaim 1 wherein the noise signal includes at least CTB noise and humnoise, and further comprising step e) of processing the noise signalestimate to obtain a measurement of CTB noise by obtaining a noisemeasurement from said noise signal estimate and substantially reducing aportion of the noise measurement that is attributable to hum noise. 3.The method of claim 2 wherein step e) further comprises obtaining thenoise measurement by first generating a frequency response of the noisesignal estimate, wherein the frequency response comprises a plurality offrequency bins, each representative of an energy of the noise signalcomponent within a predetermined frequency band.
 4. The method of claim3 wherein step e) further comprises performing a discrete Fouriertransform to generate the frequency response of the noise signalestimate.
 5. The method of claim 3 wherein step e) further comprisesremoving at least one frequency bin in which substantially all of theportion of the noise measurement that is attributable to hum noise islocated in order to substantially reduce the portion of the noisemeasurement that is attributable to hum noise.
 6. The method of claim 3wherein step e) further comprises employing a digital filter tosubstantially reduce the hum noise in the noise signal estimate beforeobtaining the noise measurement.
 7. A method of determining noise in atelevision signal, the television signal comprising a baseband signalmodulated onto the carrier signal, the method comprising: a) obtaining adigital signal segment comprising a baseband component, said basebandcomponent further comprising a noise signal component and an informationsignal component, the information signal component having asubstantially predetermined signal pattern; b) obtaining an estimate ofthe information signal and subtracting the estimate from the basebandsignal, thereby producing a noise signal estimate; c) obtaining a noisemeasurement by first generating a frequency response of the noise signalestimate, wherein the frequency response comprises a plurality offrequency bins, each representative of an energy of the noise signalcomponent within a predetermined frequency band; d) employing a digitalfilter to substantially reduce the hum noise in the noise signalestimate; and e) processing the frequency response to obtain ameasurement of one of CTB noise or CSO noise.
 8. The method of claim 7wherein step e) further comprises: processing the noise signal estimateto obtain a first measurement of CTB noise and snow noise by obtaining anoise measurement for a range of frequencies in the noise signalestimate that correspond to the frequencies in which CTB noise islocated; and processing the first measurement by substantially reducinga portion of the noise measurement that is attributable to snow noise.9. The method of claim 8 wherein step e) further comprises: determininga snow noise measurement using one or more frequency bins correspondingto a frequency band in which substantially only snow noise is present;and using the determined snow noise measurement to subtract the portionof the first measurement that is attributable to snow noise.